Position location with low tolerance oscillator

ABSTRACT

The present invention is a novel and improved method and apparatus for performing position location in wireless communications system. One embodiment of the invention comprises a method of performing position location in a wireless subscriber unit having a local oscillator, including the steps of receiving a position location request, acquiring a timing signal when a sufficient period of time has elapsed since the local oscillator has been corrected and correcting said local oscillator using a correction signal based on said timing signal, substantially freezing the correction signal, performing a position location procedure using the local oscillator with the correction signal applied, and ending said position location procedure.

BACKGROUND OF THE INVENTION

I. Field of the Invention

The present invention relates to position location. More particularly,the present invention relates to a novel and improved method andapparatus for performing position location in wireless communicationssystem.

II. Description of the Related Art

Both government regulation and consumer demand have driven the demandfor position location functionality in cellular telephones. The globalpositioning system (GPS) is currently available for performing positionlocation using a GPS receiver in conjunction with a set of earthorbiting satellites. It is therefore desirable to introduce GPSfunctionality into a cellular telephone.

Cellular telephones, however, are extremely sensitive to cost, weightand power consumption considerations. Thus, simply adding additionalcircuitry for performing GPS location is an unsatisfactory solution forproviding position location functionality in a cellular telephone. Thus,the present invention is directed to providing GPS functionality in acellular telephone system with a minimum of additional hardware, costand power consumption.

SUMMARY OF THE INVENTION

The present invention is a novel and improved method and apparatus forperforming position location in wireless communications system. Oneembodiment of the invention comprises a method of performing positionlocation in a wireless subscriber unit having a local oscillator,including the steps of receiving a position location request, acquiringa timing signal when a sufficient period of time has elapsed since thelocal oscillator has been corrected and correcting said local oscillatorusing a correction signal based on said timing signal, substantiallyfreezing the correction signal, performing a position location procedureusing the local oscillator with the correction signal applied, andending said position location procedure.

BRIEF DESCRIPTION OF THE DRAWINGS

The features, objects, and advantages of the present invention willbecome more apparent from the detailed description set forth below whentaken in conjunction with the drawings in which like referencecharacters identify correspondingly throughout and wherein:

FIG. 1 is a block diagram of the Global Positioning System (GPS)waveform generator;

FIG. 2 is a highly simplified block diagram of a cellular telephonesystem configured in accordance with the use of present invention;

FIG. 3 is a block diagram of a receiver configured in accordance withone embodiment of the invention;

FIG. 4 is another block diagram of the receiver depicted in FIG. 3;

FIG. 5 is a receiver configured in accordance with an alternativeembodiment of the invention;

FIG. 6 is a flow chart of the steps performed during a position locationoperation;

FIG. 7 is a block diagram of a DSP configured in accordance with oneembodiment of the invention;

FIG. 8 is a flow chart illustrating the steps performed during a searchperformed in accordance with one embodiment of the invention;

FIG. 9 is a time line illustrating the phases over which fine and coarsesearches are performed in one embodiment of the invention;

FIG. 10 is a time line of the search process when performed inaccordance with one embodiment of the invention;

FIG. 11 is a diagram of search space.

FIG. 12 is a block diagram of a receiver in accordance with anotherembodiment of the invention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

A novel and improved method and apparatus for performing positionlocation in wireless communications system is described. The exemplaryembodiment is described in the context of the digital cellular telephonesystem. While use within this context is advantageous, differentembodiments of the invention may be incorporated in differentenvironments or configurations. In general, the various systemsdescribed herein may be formed using software controlled processors,integrated circuits, or discreet logic, however, implementation in anintegrated circuit is preferred. The data, instructions, commands,information, signals, symbols and chips that may be referencedthroughout the application are advantageously represented by voltages,currents, electromagnetic waves, magnetic fields or particles, opticalfields or particles, or a combination thereof. Additionally, the blocksshown in each block diagram may represent hardware or method steps.

FIG. 1 is a block diagram of the Global Positioning System (GPS)waveform generator. The circle with a plus sign designates modulo-2addition. In general, the GPS constellation consists of 24 satellites:21 space vehicles (SVs) used for navigation and 3 spares. Each SVcontains a clock that is synchronized to GPS time by monitoring groundstations. To determine a position and time, a GPS receiver processes thesignals received from several satellites. At least 4 satellites must beused to solve for the 4 unknowns (x, y, z, time).

Each SV transmits 2 microwave carriers: the 1575.42 MHz L1 carrier,which carries the signals used for Standard Positioning Service (SPS),and the 1227.60 MHz L2 carrier, which carries signals needed for PrecisePositioning Service (PPS). PPS is used by governmental agencies andallows a higher degree of accuracy in positioning.

The L1 carrier is modulated by the Coarse Acquisition (C/A) code, a1023-chip pseudorandom code transmitted at 1.023 Mcps that is used forcivil position location services. (The Coarse Acquisition code shouldnot be confused with the coarse and fine acquisitions described herein,which both involve the use of the C/A codes.) Each satellite has its ownC/A code that repeats every 1 ms. The P code, which is used for PPS, isa 10.23 MHz code that is 267 days in length. The P code appears on bothcarriers but is 90 degrees out of phase with the C/A code on the L1carrier. The 50 Hz navigation message, which is exclusive-ORed with boththe C/A code and P code before carrier modulation, provides systeminformation such as satellite orbits and clock corrections.

Each satellite has a different C/A code that belongs to a family ofcodes called Gold codes. Gold codes are used because thecross-correlation between them is small. The C/A code is generated usingtwo 10-stage shift registers. A G1 generator uses the polynomial1+X³+X¹⁰, while the a G2 generator uses the polynomial1+X²+X³+X⁶+X⁸+X⁹+X¹⁰. The C/A code is generated by exclusive ORing theoutput of the G1 shift register with 2 bits of the G2 shift register.

FIG. 2 is a highly simplified block diagram of a cellular telephonesystem configured in accordance with the use of the disclosed method andapparatus. Mobile telephones 10 are located among base stations 12,which are coupled to base station controller (BSC) 14. Mobile switchingcenter MSC 16 connects BSC 14 to the public switch telephone network(PSTN). During operation, some mobile telephones are conductingtelephone calls by interfacing with base stations 12 while others are instandby mode.

As described in copending U.S. patent application Ser. No. 09/040,501now U.S. Pat. No. 6,081,229 entitled “SYSTEM AND METHOD FOR DETERMININGTHE POSITION OF A WIRELESS CDMA TRANCEIVER” assigned to the assignee ofthe present invention and incorporated herein by reference, positionlocation is facilitated by the transmission of a position requestmessage containing “aiding information” that allows the mobile telephoneto quickly acquire the GPS signal. This information includes the IDnumber of the SV (SV ID), the estimated code phase, the search windowsize around the estimate code phase, and the estimated frequencyDoppler. Using this information, the mobile unit can acquire the GPSsignals and determine its location more quickly.

In response to the aiding message, the mobile unit tunes to the GPSfrequency and begins correlating the received signal with its locallygenerated C/A sequences for the SVs indicated by the base station. Ituses the aiding information to narrow the search space and compensatefor Doppler effects, and obtains pseudo-ranges for each satellite usingtime correlation. Note that these pseudo-ranges are based on mobile unittime (referenced from the CDMA receiver's combiner system time counter),which is a delayed version of GPS time.

Once this information is calculated, the mobile unit sends thepseudo-ranges for each satellite (preferably to ⅛ chip resolution) andthe time the measurements were taken to the base station. The mobileunit then retunes to CDMA to continue the call.

Upon, receipt of the information, the BSC uses the one-way delayestimate to converts the pseudo-ranges from mobile unit time to basestation time and computes the estimated position of the mobile unit bysolving for the intersection of several spheres.

Another parameter provided by the aiding message is the frequencyDoppler or Doppler offset. The Doppler effect manifests as an apparentchange in the frequency of a received signal due to a relative velocitybetween the transmitter and receiver. The effect of the Doppler on thecarrier is referred to as frequency Doppler, while the effect on thebaseband signal is referred to as code Doppler.

In the GPS case, frequency Doppler changes the received carrierfrequency so the effect is the same as demodulating with a carrieroffset. Since the base station's GPS receiver is actively tracking thedesired satellite, it knows the frequency Doppler due to satellitemovement. Moreover, the satellite is so far away from the base stationand the mobile unit that the Doppler seen by the mobile unit iseffectively the same as the Doppler seen by the base station. In oneembodiment of the invention, to correct for the frequency Doppler value,the mobile unit uses a rotator in the receiver. The frequency Dopplerranges from −4500 Hz to +4500 Hz, and the rate of change is on the orderof 1 Hz/s.

The effect of the code Doppler is to change the 1.023 Mhz chip rate,which effectively compresses or expands the width of the received C/Acode chips. In one embodiment of the invention, the mobile unit correctfor code Doppler by multiplying the frequency Doppler by the ratio1.023/1575.42. The mobile unit can then correct for code Doppler overtime by slewing (introducing delay into) the phase of the received IQsamples in {fraction (1/16)} chip increments as necessary.

FIG. 3 is a block diagram of the receiver portion of a cellulartelephone (wireless subscriber unit) configured in accordance with oneembodiment of the invention. The received waveform 100 is modeled as theC/A signal c(n) modulated with a carrier at frequency w_(c)+w_(d), wherew_(c) is the nominal carrier frequency 1575.42 MHz, and w_(d) is theDoppler frequency created by satellite movement. The Doppler frequencyranges from 0 when the satellite is directly overhead, to about 4.5 kHzin the worst case. The receiver analog section can be modeled asdemodulation with a carrier at frequency w_(r) and random phase θ,followed by low pass filtering.

The resulting baseband signal is passed through an A/D converter (notshown) to produce digital I and Q samples, which are stored so that theymay be repeatedly searched. The samples are generated at two times theC/A code chip rate (chip×2) which is a lower resolution than necessaryto perform the fine search algorithm, but which allows 18 ms of sampledata to be stored in a reasonable amount of memory. In general, it isdesirable to perform the searching over something greater than 10 ms inorder to allow acquisition in most environmental conditions, with 18 msbeing a preferred integration period. These environmental conditionsinclude being inside or not having a direct view to the satellite.

During operation, the samples are first rotated by rotator 102 tocorrect for the Doppler frequency offset. The rotated I and Q samplesare correlated with various offsets of the satellite's C/A sequence andthe resulting products are coherently integrated over Nc chips byintegrators 104. The coherent integration sums are squared and addedtogether to remove the effect of the unknown phase offset θ. To augmentthe hypothesis test for a particular offset, several coherent intervalsare non-coherently combined. This despreading is performed repeatedly atvarious time offsets to find the time offset of the satellite signal.Rotator 102 removes the frequency Doppler created by satellite movement.It uses the Doppler frequency specified by the base station (preferablyquantized to 10 Hz intervals) and rotates the I and Q samples to removethe frequency offset.

In one embodiment of the invention, the rotation is continuous only overthe coherent integration window. That is, the rotator stops in betweencoherent integration periods of, for example, 1 ms. Any resulting phasedifference is eliminated by the square and sum.

FIG. 4 is another block diagram of a receiver configured in accordancewith one embodiment of the invention, where the rotator portion of thereceiver is depicted in greater detail.

FIG. 5 is a receiver configured in accordance with an alternativeembodiment of the invention. This internal embodiment of the inventiontakes advantage of the ability to stop the rotator between coherentintegration periods by rotating the locally generated C/A sequenceinstead of the input samples.

As shown, the C/A sequence c(n) are rotated by application to thesinusoids sin(W_(d)nT_(c)) and cos(W_(d)nT_(c)) and then stored. Therotation of the C/A sequence only needs to be done once for eachsatellite. Thus, rotating the C/A sequence reduces the amount ofcomputation required. It also saves memory in the DSP used to performthis computation in one embodiment of the invention.

Another significant impairment that degrades the performance of theposition location algorithm is the frequency error in the mobile unitsinternal clock. It is this frequency error which drives the use of shortcoherent integration times on the order of 1 ms. It is preferable toperform coherent integration over longer time periods.

In an exemplary configuration, the mobile's free running (internal)local oscillator clock is a 19.68 MHz crystal that has a frequencytolerance of +/−5 ppm. This can cause large errors on the order of+/±7500 Hz. This clock is used to generate the carriers used fordemodulation of the GPS signals, so the clock error will add to thesignal acquisition time. Because the time available to search is verysmall, error of this magnitude due to the frequency tolerance are nottolerable and must be greatly reduced.

To allow longer coherent integration times, in one embodiment of theinvention, the CDMA receiver corrects for local oscillator error byusing timing acquired from the CDMA pilot, or any other source of timinginformation available such as a GSM (Global System for MobileCommunications) pilot, a TDMA (Time Division Multiple Access) pilot.This produces a control signal that is used to tune the local oscillatorclock to 19.68 MHz as closely as possible. The control signal applied tothe local oscillator dock is frozen when the RF unit switches from CDMAto GPS.

Even after the correction is performed using the timing information fromthe bases station (or other source), however, some additional clockerror remains. In one embodiment of the invention, the resultingfrequency uncertainty after correction is +/−100 Hz. This remainingerror still reduces the performance of the receiver, and in generalprevents longer coherent integration times. In one embodiment of theinvention, the remaining error simply avoided by performing non-coherentintegration for duration of more than 1 ms which reduces performance.

As also shown in FIG. 1, the 50 Hz NAV/system data is also modulatedonto the L1 carrier. If a data transition (0 to 1 or 1 to 0) occursbetween the two halves of a coherent integration window, the resultingcoherent integration sum will be zero because the two halves will canceleach other out. This effectively reduces the number of non-coherentaccumulations by one in the worst case. Although the data boundaries ofall the satellites are synchronized, they do not arrive at the mobileunit simultaneously because of the differences in path delay. This pathdelay effectively randomizes the received data phase.

In one embodiment of the solution to the invention, the problem ofdifferent data phases on different signals is to include the data phasein the aiding information sent from the base station to the mobile unit.Since the base station is demodulating the 50 Hz data, it knows when thedata transitions occur for each satellite. By using knowledge of theone-way delay, the base station can encode the data phase in, forexample, 5 bits (per satellite) by indicating which one millisecondinterval (out of 20) the data transition occurs on.

If the coherent integration window straddles the 50 Hz data boundary thecoherent integration is divided into two (2) sections. One sectionpreceding the data boundary and one section following the data boundary.For example, if En1 is the coherent integration sum over the windowpreceding the data boundary the first half of this window and En2 is thecoherent integration sum over the window following the data boundary,the mobile unit then selects the maximum (in magnitude) of (En1+En2) (incase the data stayed the same) and (En1−En2) (in case the data changed)to account for the phase change. The mobile unit also has the option ofperforming non-coherent combining of the two halves over this datawindow or avoiding this data window completely.

In an alternative embodiment of the invention, the mobile unit attemptsto find the data transitions without the aiding information from thebase station by comparing the magnitude squared of the sum anddifference in 1 ms coherent integration.

In one embodiment of the invention, a firmware-based DSP (Digital SignalProcessor) approach is used to perform the GPS processing. The DSPreceives I and Q samples at a chip×2 (2.046 MHz) or chip×8 (8.184 MHz)rate, and stores a snapshot of 4-bit I and Q samples in its internalRAM.

In the exemplary embodiment, the DSP generates the C/A sequence,performs rotation to eliminate frequency Doppler, and correlates overthe search window provided by the base station for each of thesatellites. The DSP performs coherent integration and non-coherentcombining and slews an IQ sample decimator as necessary to compensatefor code Doppler.

To save computation and memory, the initial search is performed using ½chip resolution and a finer search to obtain ⅛ chip (higher) resolutionis performed around the best index (or indexes). System time ismaintained by counting hardware-generated 1 ms interrupts (derived fromlocal oscillator).

Additionally, in one embodiment of the invention, the fine search isperformed by accumulating the chip×8 samples (higher resolution) overthe duration of one chip at various chip×8 offsets. The correlationcodes are applied to the accumulated values yielding correlation valuesthat vary with the particular chip×8 offset. This allows the code offsetto be determined with chip×8 resolution.

FIG. 6 is a flow chart illustrating the steps performed to correct forthe local oscillator error during a position location procedure whenperformed in accordance with one embodiment of the invention. At step500, it is determined whether the local oscillator has been correctedrecently. If not, then the pilot is acquired from the base station, anderror of the local oscillator is determined by comparing to the pilottiming at step 502 and a correction signal generated based on thaterror.

The flow then leads to step 504, where the correction signal is frozenat the current value. At step 506, enters GPS mode and performs theposition location using the corrected clock. Once the position locationhas been performed, the mobile unit leaves GPS mode at step 508.

FIG. 7 is an illustration of a DSP receiver system configured inaccordance with one embodiment of the invention. The DSP performs theentire searching operation with minimal additional hardware. DSP core308, modem 306, interface unit 300, ROM 302 and Memory (RAM) 304 arecoupled via bus 306. Interface unit 300 receives RF samples from an RFunit (not shown) and provides the samples to RAM 304. The RF samples canbe stored at coarse resolution or fine resolution. The DSP core 308processes the samples stored in memory using instruction stored in ROM302 as well as in memory 304. Memory 304 may have multiple “banks” someof which store samples and some of which store instructions. Modem 700performs CDMA processing during normal mode.

FIG. 8 is a flow chart of the steps performed during a position locationoperation. A position location operation begins when the aidingmessaging is received, and the RF systems is switched to GPS frequenciesat step 600. When the RF is switched to receive GPS, the frequencytracking loop is fixed. The DSP receives aiding information from thephone microprocessor and sorts the satellites by Doppler magnitude.

At step 602, the coarse search data is stored within the DSP RAM. TheDSP receives a few hundred microseconds of input data to set an Rx AGC.The DSP records the system time and begins storing an 18 ms window (DSPmemory limitation) of chip×2 IQ data in its internal RAM. A contiguouswindow of data is used to mitigate the effects of code Doppler.

Once the data is stored, a coarse search is performed at step 604. TheDSP begins the coarse (chip×2 resolution) search. For each satellite,the DSP generates the C/A code, rotates the code based on the frequencyDoppler, and correlates over the search window specified by the basestation, via repeated application of the C/A code to the stored coarsesearch data. Satellites are processed over the same 18 ms data windowand the best chip×2 hypothesis that exceeds a threshold is obtained foreach satellite. Although a 2 ms coherent integration time (with 9non-coherent integrations) is used in one embodiment of the invention,longer coherent integration times can be used (for example 18 ms),although preferably where additional adjustments are made as describedbelow.

Once the coarse search is performed, a fine search is conducted, at step606. Before beginning the fine search, the DSP computes the rotated C/Acode for each of the satellites. This allows the DSP to process the finesearch in real-time. In performing the fine (chip×8 resolution) search,the satellites are processed one at a time over different data.

The DSP first slews the decimator to compensate for code Doppler for thegiven satellite(s). It also resets the Rx AGC value while waiting forthe next 1 ms boundary before storing a 1 ms coherent integration windowof chip×8 samples.

The DSP processes 5 contiguous chip×8 resolution hypotheses on this 1 mscoherent integration window, where the center hypothesis is the besthypothesis obtained in the coarse search. After processing the next 1 mswindow, the results are combined coherently and this 2 ms sum iscombined non-coherently for all Nn iterations.

This step (starting from slewing the decimator) is repeated on the samedata for the next satellite until all the satellites have beenprocessed. If the code Doppler for 2 satellites is similar in magnitude,it may be possible to process both satellites over the same data toreduce the number of required data sets. In the worst case, 8 sets of 2*Nn data windows of 1 ms are used for the fine search.

Finally, at step 608, the results are reported to the microprocessor andthe vocoder process is restarted within the DSP so that the call cancontinue. The DSP reports pseudoranges to the microprocessor, whichforwards them to the base station. After the microprocessor redownloadsthe vocoder program code into the DSP memory, the DSP clears its datamemory and restarts the vocoder.

FIG. 9 is a diagram illustrating the fine search performed after thecoarse search. After isolating the best chip×2 phase in the coarsesearch, the DSP performs a fine search around this phase to gain chip×8resolution.

The 5 phases to compare in the fine search are shown enclosed by arectangle. The best chip×2 phase is evaluated again so that comparisonscan be made over the same set of data. This also allows the coarsesearch and fine search to use different integration times. The finesearch is performed separately for each satellite because each satellitemay have a different value for code Doppler.

FIG. 10 provides a time line of the search process when performed inaccordance with one embodiment of the invention. The overall processingtime (coarse+fine search) is performed in about 1.324 seconds in oneembodiment of the invention, which does interrupt the call, but stillallows the call to continue once the search is performed. The totalsearch time of 1.324 seconds is an upper bound, because it assumes thatthe DSP needs to search all 8 satellites and each satellite has a searchwindow of 68 chips. The probability that the entire 1.324 seconds willbe necessary is small, however, due to the geometry of the satelliteorbits.

During the first 18 ms 80, IQ sample data is collected at the GPSfrequency. During the period 82, a coarse search is performed internallywhich could last up to 1.13 seconds, but which will probably terminateearly when the satellite signals are identified. Once the coarse searchis performed, the C/A codes are computed during time period 84, whichtakes 24 ms. During time periods 86 the slew value is adjusted for codeDoppler and the Rx AGC is further adjusted. During time periods 88, finesearches are performed on the IQ data samples, with continuousadjustment performed during time periods 86. The use of 18 msintegration times allows code Doppler to be neglected because thereceived C/A code phase will be shifted by less than {fraction (1/16)}of a chip. Up to eight sequences of adjustments and fine searches areperformed for the up to eight satellites, at which time the positionlocation procedure is complete.

Additionally, in some embodiments of the invention, the phone continuesto transmit reverse link frames to the base station while the positionlocation procedure is performed. These frames may contain nullinformation simply to allow the base station to remain synchronized withthe subscriber unit, or the frames may contain additional informationsuch as power control commands or information request. The transmissionof these frames is preferably performed when GPS samples are not beinggathered when the RF circuitry is available, or while the GPS samplesare gathered if sufficient RF circuitry is available.

Although the use of 18 ms integration time avoids the effects of codeDoppler, the transmission of data over the GPS signals at 50 Hz rate cancause problems if a data change occurs within the 18 ms processing span(as described above). The data change causes the phase of the signal toshift. The 50 Hz data boundaries occur at different places for eachsatellite. The phase of the 50 Hz transitions for each satellite havebeen effectively randomized by the varying path lengths from eachsatellite to the phone.

In the worst case, if the data bit was inverted in the middle of acoherent integration interval, the coherent integration could becompletely wiped out. For this reason, in an alternative embodiment ofthe invention, the base station must communicate the data transitionboundaries for each satellite to the phone (also described above).Preferably, the data transmission boundary is also included in theaiding message transmitted from the base station (such as in a set offive bit messages indicating the millisecond interval during which thetransition occurs for each satellite). The phone uses this boundary tosplit the coherent integration interval for each satellite into 2 piecesand decide whether to add or subtract the coherent integration sums inthese 2 intervals. Thus, by also including the data boundary of each GPSsignal, the reliability of the location procedure is increased.

In the exemplary embodiment of the invention, any frequency uncertaintycreates a loss in Ec/Nt that increases with the coherent integrationtime. For example, uncertainty of +/−100 Hz, the loss in Ec/Nt increasesrapidly as the coherent integration time is increased, as shown in TableI.

TABLE I Nc Loss in Ec/Nt 1023 (1 ms) 0.14 dB 2046 (2 ms) 0.58 dB 4092 (4ms) 2.42 dB 6138 (6 ms) 5.94 dB 8184 (8 ms) 12.6 dB

As also noted above, there is always some unknown frequency offset ofthe local oscillator in the mobile unit. It is this unknown frequencyoffset that prevents longer coherent despreading and integration frombeing performed. Longer coherent would improve processing if the effectsof the unknown frequency offset could be reduced.

In one embodiment of the invention, this unknown frequency offset isaccounted for by expanding the search space to 2 dimensions to includefrequency searches. For each hypothesis, several frequency searches areperformed, where each frequency search assumes the frequency offset is aknown value. By spacing the frequency offsets, one can reduce thefrequency uncertainty to an arbitrarily small value at the expense ofadded computation and memory. For example, if 5 frequency hypotheses areused, the resulting search space is shown in FIG. 10.

For a +/−10 Hz frequency uncertainty, which is the typically operatingspecification of a mobile unit, this configuration reduces the maximumfrequency offset to 20 Hz (one hypothesis must be within 20 Hz of theactual frequency offset). With a 20 ms coherent integration time, theloss in Ec/Nt with a 20 Hz frequency offset is 2.42 dB. By doubling thenumber of frequency hypotheses to 10, the frequency uncertainty can bereduced to 10 Hz, which causes an Ec/Nt loss of 0.58 dB. However, addingadditional hypotheses widens the search space, which increases both thecomputation and memory requirements.

One embodiment of the invention computes the frequency hypothesis bylumping the frequency offset in with the frequency Doppler, andcomputing a new rotated PN code for each frequency hypothesis. However,this makes the number of frequency hypotheses a multiplicative factor inthe total computation: 5 frequency hypotheses would mean 5 times as muchcomputation.

Alternatively, since the frequency uncertainty is small compared to thefrequency Doppler, the rotation phase can be considered to be constantover a 1 ms interval (8% of a period for an 80 Hz hypothesis) in anotherembodiment of the invention. Therefore, by dividing the coherentintegration interval up into 1 ms subintervals, the integration sums ofthe subintervals are rotated to reduce the added computations needed tocompute the frequency searches by three orders of magnitude. The resultis that longer coherent despreading can be performed, and performanceimproved.

FIG. 12. is a block diagram of a receiver configured in accordance withthe use of longer coherent despreading approach. The first set ofmultipliers 50 compensates for the frequency Doppler by correlating theIQ samples with a rotated C/A code. This is equivalent to rotating theIQ samples before correlation with the unmodified C/A code. Since thefrequency Doppler can be as large as 4500 Hz, the rotation is applied toevery chip. After coherently integrating over a 1 ms interval (1023chips) using accumulators 52, the second set of multipliers rotates the1 ms integration sums (Σ_(I) and Σ_(Q)) to implement the frequencyhypothesis. The rotated sums are then added over the whole coherentintegration interval.

Recall that the frequency Doppler rotation was only computed on 1023chips to save memory and computation. For coherent integration timeslonger than 1 ms, each coherent integration sum are multiplied by aphase offset to make the phase of the rotation continuous over time. Toshow this mathematically, the 1 ms coherent integration sum withfrequency Doppler rotation can be expressed as:$S_{1} = {\sum\limits_{n = 1}^{1023}{\left\lbrack {{I(n)} + {{jQ}(n)}} \right\rbrack \quad {c(n)}^{{- j}\quad w_{d}{nT}_{c}}}}$with  Σ_(I) = Re{S₁}  and  Σ_(Q) = Im{S₁}

where I(n) and Q(n) are the input samples received on the I and Qchannels respectively, c(n) is the unrotated C/A code, w_(d) is thefrequency Doppler, and T_(c) is the chip interval (0.9775 us). A 2 mscoherent integration sum can be expressed as: $\begin{matrix}{{S\left( {2{ms}} \right)} = \quad {\sum\limits_{n = 1}^{2046}{\left\lbrack {{I(n)} + {{jQ}(n)}} \right\rbrack \quad {c(n)}^{{- j}\quad w_{d}{nT}_{c}}}}} \\{= \quad {{\sum\limits_{n = 1}^{1023}{\left\lbrack {{I(n)} + {{jQ}(n)}} \right\rbrack \quad {c(n)}^{{- j}\quad w_{d}{nT}_{c}}}} +}} \\{\quad {^{{- j}\quad {w_{d}{(1023)}}T_{c}}{\sum\limits_{n = 1}^{1023}\left\lbrack {{I\left( {n + 1023} \right)} +} \right.}}} \\{{\quad \left. {{jQ}\left( {n + 1023} \right)} \right\rbrack}\quad {c(n)}^{{- j}\quad w_{d}{nT}_{c}}} \\{= \quad {S_{1} + {^{{- j}\quad {w_{d}{(1023)}}T_{c}}S_{2}}}}\end{matrix}$

Here, S₁ is the first 1 ms integration sum and S₂ is the second 1 msintegration sum computed using the same rotated C/A values that wereused to compute S₁. The term e^(−jwd(1023)Tc) is the phase offset thatcompensates for using the same rotated values. Similarly, a 3 mscoherent integration sum can be expressed as

S(3 ms)=S ₁ +e ^(−jw) ^(_(d)) ^((1023)T) ^(_(c)) S ₂ +e ^(−jw) ^(_(d))^((2046)T) ^(_(c)) S ₃

So to extend the integration time while using the same 1023-elementrotated C/A sequence, the (n+1) 1 ms integration sum should bemultiplied by e^(−jwdn(1ms)) before being added to the whole sum. Sincethis is a rotation of 1 ms integration sums, we can combine thisoperation with the frequency search to avoid having to perform 2rotations. That is, since

e ^(−jw) ^(_(d)) ^(n(1ms)) e ^(−jw) ^(_(h)) ^(n(1ms)) =e ^(−j(w) ^(_(d))^(+w) ^(_(h)) ^()n(1ms))

we can multiply the (n+1)th 1 ms integration sum by e^(−j(wd+wh)n(1ms))to search a frequency hypothesis and account for the frequency Dopplerphase offset.

Note that the frequency search can be reduced after acquiring onesatellite, because the frequency uncertainty is not dependent on thesatellite. A much finer frequency search can be performed if a longercoherent integration is desired.

In the exemplary embodiment of the invention, the fine search isperformed in similar manner the coarse search with 2 differences. First,the integration intervals are always added coherently instead ofsquaring and adding noncoherently. Second, the rotation to remove thefrequency uncertainty (which should be known after the coarse search) iscombined with the frequency Doppler phase offset and used to rotate the1 ms coherent integration intervals before adding them together.

In an alternative embodiment of the invention, the coherent integrationwindow of chip×2 data is integrated for integration times longer than 18ms. This embodiment is useful were additional memory is available. Forcoherent integrations longer than 18 ms, the 50 Hz data boundaries aretreated the same as with shorter integration periods. The base stationindicates where the boundaries are for each satellite and the DSPdecides whether to add or subtract the sum of 20 1 ms coherentintegration intervals to or from its running sum.

However, because the product of the frequency uncertainty and theintegration time constant affects the loss in Ec/Nt, the frequencyuncertainty must be reduced to very small levels for long coherentintegration intervals. Since a 20 ms integration with a 20 Hz frequencyuncertainty resulted in a loss in Ec/Nt of 2.42 dB, maintaining the sameloss with an integration time of 400 ms requires that the frequencyuncertainty be reduced to 1 Hz. To correct for this problem, thefrequency uncertainty is reduced down to 1 Hz in a hierarchical manner.For example, a first frequency search reduces the uncertainty from 100Hz to 20 Hz, a second search reduces the uncertainty to 4 Hz, and athird search reduces the uncertainty to 1 Hz. The frequency search willalso compensate for errors in the frequency Doppler obtained from thebase station.

Additionally, to perform longer integrations only satellites withsimilar Doppler are searched over the same data for long integrationtimes, since the code Doppler is different for each satellite. The DSPcomputes how long it takes to slip {fraction (1/16)} of a chip and slewsthe decimator as it collects a coherent integration data window.Additionally, multiple data windows are taken in this embodiment.

Thus, a method and apparatus for performing position location inwireless communications system has been described. The previousdescription of the preferred embodiments is provided to enable anyperson skilled in the art to make or use the present invention. Thevarious modifications to these embodiments will be readily apparent tothose skilled in the art, and the generic principles defined herein maybe applied to other embodiments without the use of the inventivefaculty. Thus, the present invention is not intended to be limited tothe embodiments shown herein but is to be accorded the widest scopeconsistent with the principles and novel features disclosed herein.

What is claimed is:
 1. A method of performing position location in awireless subscriber unit having a local oscillator, comprising the stepsof: receiving a position location request; acquiring a timing signal aperiod of time after the local oscillator had been corrected andcorrecting said local oscillator using a correction signal based on saidtiming signal; substantially freezing the correction signal; performinga position location procedure using the local oscillator with thecorrection signal applied; and ending said position location procedure.2. The method as set forth in claim 1 wherein said timing signal is apilot signal of a wireless communication system.
 3. The method as setforth in claim 2 wherein said pilot signal is a CDMA pilot signal. 4.The method as set forth in claim 2 wherein said pilot signal is a GSMpilot signal.
 5. The method as set forth in claim 2 wherein said pilotsignal is a TDMA pilot signal.
 6. The method of claim 1 furthercomprising the step of entering a communications mode after performingposition location procedure.